***Properties Analysis of DWDM and CWDM GPON***


A PON takes advantage of wavelength division multiplexing (WDM), using one wavelength for downstream traffic and another for upstream traffic on a single non-zero dispersion-shifted fiber (ITU-T G.652). BPON, EPON, GEPON, and GPON have the same basic wavelength plan and use the 1490 nano meter (nm) wavelength for downstream traffic and 1,310 nm wavelength for upstream traffic. 1,550 nm is reserved for optional overlay services, typically RF (analog) video.

As with bit rate, the standards describe several optical budgets, most common is 28 dB of loss budget for both BPON and GPON, but products have been announced using less expensive optics as well. 28 dB corresponds to about 20 km with a 32-way split. Forward error correction (FEC) may provide another 2–3 dB of loss budget on GPON systems. As optics improve, the 28 dB budget will likely increase. Although both the GPON and EPON protocols permit large split ratios (up to 128 subscribers for GPON, up to 32,768 for EPON), in practice most PONs are deployed with a split ratio of 1×32 or smaller.

The access network, also known as the “first-mile network,” connects the service provider central offices (COs) to businesses and residential subscribers. This network is also referred to in the literature as the subscriber access network, or the local loop. The bandwidth demand in the access network has been increasing rapidly over the past several years. Residential subscribers demand first-mile access solutions that have high bandwidth and offer media-rich services. Similarly, corporate users demand broadband infrastructure through which they can connect their local-area networks to the Internet backbone.

The predominant broadband access solutions deployed today are the digital subscriber

line (DSL) and community antenna television (CATV) (cable TV) based networks. However, both of these technologies have limitations because they are based on infrastructure that was originally built for carrying voice and analog TV signals, respectively; but their retrofitted versions to carry data are not optimal. Currently deployed blends of asymmetric DSL (ADSL) technologies provide 1.5 Mbits/s of downstream bandwidth and 128 Kbits/s of upstream bandwidth at best. Moreover, the distance of any DSL subscriber to a CO must be less than 18000 ft because of signal distortions. Although variations of DSL such as very-high-bit-rate DSL (VDSL), which can support up to 50 Mbits/s of downstream bandwidth, are gradually emerging, these technologies have much more severe distance limitations.

For example, the maximum distance over which VDSL can be supported is limited to 1500 ft. CATV networks provide Internet services by dedicating some radio frequency (RF) channels in a coaxial cable for data. However, CATV networks are mainly built for delivering broadcast services, so they don’t fit well for the bidirectional communication model of a data network. At high load, the network’s performance is usually frustrating to end users.

Passive optical networks (PONs) have evolved to provide much higher bandwidth in the access network. A PON is a point-to-multipoint optical network, where an optical line terminal (OLT) at the CO is connected to many optical network units (ONUs) at remote nodes through one or multiple 1:N optical splitters. The network between the OLT and the ONU is passive, i.e., it does not require any power supply.

PONs use a single wavelength in each of the two directions—downstream (CO to end users) and upstream (end users to CO)—and the wavelengths are multiplexed on the same fiber through coarse WDM (CWDM). For example, the Ethernet PON (EPON) uses 1490 nm wavelength for downstream traffic and the 1310 nm wavelength for upstream traffic. Thus, the bandwidth available in a single wavelength is shared amongst all end users. Such a solution was envisaged primarily to keep the cost of the access network low and economically feasible for subscribers. Various blends of the PON have emerged in recent years: the Ethernet PON (EPON) is a relatively recent version that is standardized in the IEEE 802.3ah [1], the broadband PON (BPON) is standardized in the ITU-T G.983, and the generic framing procedure based PON (GFP PON) is standardized in the ITU-T G.984. An enhancement of the PON supports an additional downstream wavelength, which may be used to carry video and CATV services separately. Many telecom operators are considering to deploy PONs using a fiber-to-the-x (FTTx) model (where x = building (B), curb (C), home (H), premises (P), etc.) to support converged Internet protocol (IP) video, voice, and data services—defined as “triple play”—at a cheaper subscription cost than the cumulative of the above services deployed separately. PONs are in the initial stages of deployment in many parts of the world. Although the PON provides higher bandwidth than traditional copper-based access networks, there exists the need for further increasing the bandwidth of the PON by employing wavelength-division multiplexing (WDM) so that multiple wavelengths may be supported in either or both upstream and downstream directions. Such a PON is known as a WDM-PON. Interestingly, architectures for WDM-PONs have been proposed as early as the mid-1990s. However, these ideas have not been commercialized yet for many reasons: lack of an available market requiring high bandwidth, immature device technologies, and a lack of suitable network protocols and software to support the architecture. We believe that many of the above factors have been mitigated over the years, and WDM-PONs will soon be viable for commercial deployment.

GPON Triple Play Present Structure by DWDM Technology 


Traditional Triple Play Architecture
                                    Figure: Traditional Triple Play Architecture


The video signal enters to OLT through EDFA and the ISP connection for Data and Voice enter via a layer-2 switch. The OLT passed the modulated signal through fiber, optical splitter and finally from ONT’s ethernet port, the user receives the signal.

Dense Wavelength Division Multiplexing (DWDM) is an optical multiplexing technology used to increase bandwidth over existing fiber networks. DWDM works by combining and transmitting multiple signals simultaneously at different wavelengths on the same fiber. The technology creates multiple virtual fibers, thus multiplying the capacity of the physical medium. DWDM is a fiber optic transmission technique that allows the transmission of a variety of information over the optical layer. The DWDM uses dispersion-flattened fibers where the dispersion weakly depends on operating wavelength. DWDM technology is efficiently used for increasing the capacity and reliability of fiber optic communication systems. Unlike previous generation optical networks, where the information is carried by a single light beam, DWDM carves up large bandwidth of an optical fiber into many wavelength channels making spectral band use more efficient. Each of the optical carrier’s wavelength carries information individually but their spacing needs to be properly chosen to avoid inter-channel interference. DWDM can increase the information carrying capacity by about 10-100 times without the need of a new optical fiber.

DWDM systems are also bit rate and format independent and can accept any combination of interference rates on the same fiber at the same time. This technology can be applied to different areas in the telecommunications networks, that includes the backbone networks, the residential access networks and also local area networks.

Key points of DWDM

  •  DWDM optical systems require a thermoelectric cooler to stabilize the wavelength emission and absorb the power dissipated by the laser.
  • This consumes power while adding cost.
  • For short transmission distances, a ‘coarse’ wavelength grid can reduce terminal costs by eliminating the temperature control and allowing the emitted wavelengths to drift with ambient temperature changes.


DWDM System Advantages

  • Less fiber cores to transmit and receive high capacity data.
  • A single core fiber cable be could have divided into multiple channels instead of using 12 fiber cores.
  • Easy network expansion, especially for limited fiber resource; no need extra fiber but add wavelength. Low cost for expansion, because no need to replace many components such as optical amplifiers, can move to STM-64 when economics improve.
  • DWDM systems capable of longer span lengths, TDM approach using STM-64 is more costly and more susceptible to chromatic and polarization mode dispersion.

DWDM Disadvantages

  • Not cost-effective for low channels; low channel recommends CWDM
  • Complicated transmitters and receivers
  • Wide-band channel; CAPEX and OPEX high
  • The frequency domain involved in the network design and management, increases the difficulty for implementation.

Different Parameter of Calculations to compare CWDM with DWMD

 Bit Error Rate

The bit error rate (BER) is the number of erroneous bit to the total transmitted bits at the receiver that have been altered due to noise, interference and/or distortion. The modulated signals are transmitted over the optical fiber where they undergo attenuation and dispersion, have noise added to them from optical amplifiers and sustain a variety of other impairments. At the receiver, the transmitted data must be recovered with an acceptable BER. The required BER for high-speed optical communication systems today is in the range of 10-9 – 10-15, with a typical value of 10-12. A BER of 10-12 corresponds to one allowed bit error for every terabit of data transmitted on average.

Bit Error Rate of an Ideal Receiver

In principle, the demodulation process can be quite simple. Ideally, it can be viewed as “photon counting,” which is the viewpoint we will take in this section. In practice, there are various impairments that are not accounted for by this model and we discuss them in the next section.

The receiver looks for the presence or absence of light during a bit interval. If no light is seen, it infers that a 0 bit was transmitted and if any light is seen, it infers that a 1 bit was transmitted. This is called direct detection. Unfortunately, even in the absence of other forms of noise, this will not lead to an ideal error-free system because of the random nature of photon arrivals at the receiver. A light signal arriving with power P can be thought of as a stream of photons arriving at average rate P/hfc. Here, h is Planck’s constant (6.63 x 10-34 J/Hz), fc is the carrier frequency and hfc is the energy of a single photon. This stream can be thought of as a Poisson random process.

Note that our simple receiver does not make any errors when a 0 bit is transmitted. However, when a 1 bit is transmitted, the receiver may decide that a 0 bit was transmitted if no photons were received during that bit interval. If B denotes the bit rate, then the probability that n photons are received during a bit interval 1/B is given by-

Thus, the probability of not receiving any photons is  . Assuming equally likely ls and 0s, the bit error rate of this ideal receiver would be given as Let M = P/hfcB. The parameter M represents the average number of photons received during a 1 bit. Then, the bit error rate can be expressed as

This expression represents the error rate of an ideal receiver and is called the quantum limit. To get a bit error rate of 10-12, we would need an average of M = 27 photons per 1 bit.

In practice, most receivers are not ideal and their performance is not as good as that of the ideal receiver because they must contend with various other forms of noise, as we shall soon see.

Bit Error Rate of a Practical Direct Detection Receiver

The optical signal at the receiver is first photo detected to convert it into an electrical current. The main complication in recovering the transmitted bit is that in addition to the photocurrent due to the signal there are usually three others additional noise currents. The first is the thermal noise current due to the random motion of electrons that is always present at any finite temperature. The second is the shot noise current due to the random distribution of the electrons generated by the photodetection process even when the input light intensity is constant. The shot noise current, unlike the thermal noise current, is not added to the generated photocurrent but is merely a convenient representation of the variability in the generated photocurrent as a separate component. The third source of noise is the spontaneous emission due to optical amplifiers that may be used between the source and the photodetector.

The thermal noise current in a resistor R at temperature T can be modeled as a Gaussian random process with zero mean and autocorrelation function (4kBT/R)δ(τ). Here, kB is Boltzmann’s constant and has the value 1.38 x 10-23 J/0K and δ(τ) is the Dirac delta function, defined as δ(τ)=0, τ ≠ 0 and . Thus, the noise is white and in a bandwidth or frequency range Be, the thermal noise current has the variance

This value can be expressed as  where It is the parameter used to specify the current standard deviation in units of. Typical values are of the order of

The electrical bandwidth of the receiver, Be, is chosen based on the bit rate of the signal. In practice, Be varies from 1/2T to 1/T, where T is the bit period. We will also be using the parameter B0 to denote the optical bandwidth seen by the receiver. The optical bandwidth of the receiver itself is very large, but the value of B0 is usually determined by filters placed in the optical path between the transmitter and receiver.

By convention, we will measure Be in baseband units and B0  in passband units. Therefore, the minimum possible value of B0 = 2 Be, to prevent signal distortion. The photon arrivals are accurately modeled by a Poisson random process. The photocurrent can be modeled as a stream of electronic charge impulses, each generated whenever a photon arrives at the photodetector. For signal powers that are usually encountered in optical communication systems, the photocurrent can be modeled as I = + is, where is a constant current and is is a Gaussian random process with mean zero and autocorrelation  For pin diodes, . The constant current =RP, where R is the responsivity of the photodetector and P is the optical power. Here, we are assuming that the dark current, which is the photocurrent that is present in the absence of an input optical signal, is negligible. Thus, the shot noise current is also white and in a bandwidth Be  has the variance

If we denote the load resistor of the photodetector by RL , the total current in this resistor can be written as  I = + is +it ,where it has the variance –The shot noise and thermal noise currents are assumed to be independent so that, if Be is the bandwidth of the receiver, this current can be modeled as a Gaussian random process with mean  and variance

Note that both the shot noise and thermal noise variances are proportional to the bandwidth Be of the receiver. Thus, there is a trade-off between the bandwidth of a receiver and its noise performance. A receiver is usually designed so as to have just sufficient bandwidth to accommodate the desired bit rate so that its noise performance is optimized. In most practical direct detection receivers, the variance of the thermal noise component is much larger than the variance of the shot noise and determines the performance of the receiver.

Now, we will calculate the BER of a practical direct detection receiver. The receiver makes decisions as to which bit (0 or 1) was transmitted in each bit interval by sampling the photocurrent. Because of the presence of noise currents, the receiver could make a wrong decision resulting in an erroneous bit. In order to compute this bit error rate, we must understand the process by which the receiver makes a decision regarding the transmitted bit.

Consider a pin receiver. For a transmitted 1 bit, let the received optical power P = P1 and let the mean photocurrent  = I1. Then I1 = RP1 and the variance of the photocurrent is-

If P0 and I0 are the corresponding quantities for a 0 bit, I0= RP0 and the variance of the photocurrent is-

For ideal on-off-keying (OOK), P0 and I0 are zero, but this is not always the case in practice.

Let I1 and I0 denote the photocurrent sampled by the receiver during a 1 bit and a 0 bit, respectively, and let a  and  represent the corresponding noise variances. The noise signals are assumed to be Gaussian. The actual variances will depend on the type of receiver. So, the bit decision problem faced by the receiver has the following mathematical formulation. The photocurrent for a 1 bit is a sample of a Gaussian random variable with mean I1 and variance ( and similarly for the 0 bit as well). The receiver must look at this sample and decide whether the transmitted bit is a 0 or a 1. The possible probability density functions of the sampled photocurrent are sketched in Figure-

Figure -Probability density functions for the observed photocurrent.


There are many possible decision rules that the receiver can use; the receiver’s objective is to choose the one that minimizes the bit error rate. This optimum decision rule can be shown to be the one that, given the observed photocurrent I, chooses the bit (0 or 1) that was most likely to have been transmitted. Furthermore, this optimum decision rule can be implemented as follows. Compare the observed photocurrent to a decision threshold Ith . If IIth, decide that a 1 bit was transmitted; otherwise, decide that a 0 bit was transmitted.

For the case when 1 and 0 bits are equally likely (which is the only case we consider in this thesis), the threshold photocurrent is given approximately by –

This value is very close but not exactly equal to the optimal value of the threshold. The probability of error when a 1 was transmitted is the probability that I < Ith and is denoted by P[0|1]. Likewise, P[1|0] is the probability of deciding that a 1 was transmitted when actually a 0 was transmitted and is the probability that IIth. Both probabilities are indicated in above Figure.

Let Q(x) denote the probability that a zero mean, unit variance Gaussian random variable exceeds the value x. Thus-

It now follows that   and 

Using above equation, the BER is given by-

The Q function can be numerically evaluated. Let γ= Q-1 (BER). For a BER rate of   10-12, we need γ ≈ 7. For a BER rate of 10-9, γ ≈6 .

It is particularly important to have a variable threshold setting in receivers if they must operate in systems with signal-dependent noise, such as optical amplifier noise. Many high-speed receivers do incorporate such a feature. However, many of the simpler receivers do not have a variable threshold adjustment and set their threshold corresponding to the average received current level, namely, (I1 +  I0)/2. This threshold setting yields a higher bit error rate given by-

Using the above equation to evaluate the BER when the received signal powers for a 0 bit and a 1 bit and the noise statistics are known.

 Power Penalty

Power penalty is the extra power required to account for degradations due to different impairments that are present in the system. Usually each impairment results in a power penalty to the system. In the presence of an impairment, a higher signal power will be required at the receiver in order to maintain a desired bit error rate. One way to define the power penalty is as the increase in signal power required (in dB) to maintain the same bit error rate in the presence of impairments. Another way to define the power penalty is as the reduction in signal-to-noise ratio as quantified by the value of γ (the argument to the Q(.) function) due to a specific impairment. Here, we have used the latter definition since it is easier to calculate and consistent with popular usage.

Let P1 denote the optical power received during a 1 bit and P0 the power received during a 0 bit without any system impairments. The corresponding electrical currents are given by RP1 and RP0 respectively, where R  is the responsivity of the photodetector.

Let σ1 and σ0 denote the noise standard deviations during a 1 bit and a 0 bit respectively. We also assume that the noise is Gaussian. Assuming equally likely ls and 0s, the bit error rate is obtained from the above equation as-

In the presence of impairments, let denote the received powers and noise standard deviations, respectively. Assuming an optimized threshold setting, the power penalty is given by –

The case of interest is when the dominant noise component is receiver thermal noise, for which This is usually the case in unamplified direct detection pin receivers. In this case, or in any situation where the noise is independent of the signal power, the power penalty is given by-

Relative Intensity Noise (RIN)

 Relative intensity noise (RIN) is the noise of the optical intensity (or actually power), normalized to its average value. In the context of intensity noise (optical power fluctuations) of a laser, it is common to specify the relative intensity noise (RIN), which is the power noise normalized to the average power level. The optical power of the laser can be considered to be-

with an average value and a fluctuating quantity δP with zero mean value. The relative intensity noise is then that of δP divided by the average power.

Relative intensity noise can be generated from cavity vibration, fluctuations in the laser gain medium or simply from transferred intensity noise from a pump source.

RIN is the signal-dependent noise. The optical feedback from multiple reflections along the fiber path can increase the effect of RIN.

Classification of Crosstalk

Crosstalk is the general term given to the effect of other signals on the desired signal. Almost every component in a WDM system introduces crosstalk of some form or another. The components include filters, wavelength multiplexers/demultiplexers, switches, semiconductor optical amplifiers and the fiber itself (by way of nonlinearities). Two forms of crosstalk arise in WDM systems: interchannel crosstalk and intrachannel crosstalk. The first case is when the crosstalk signal is at a wavelength sufficiently different from the desired signal’s wavelength. This form of crosstalk is called interchannel crosstalk. Interchannel crosstalk can also occur through more indirect interactions. The second case is when the crosstalk signal is at the same wavelength as that of the desired signal. This form of crosstalk is called intrachannel crosstalk or also called homodyne crosstalk. Intrachannel crosstalk effects can be much more severe than interchannel crosstalk. In both cases, crosstalk results in a power penalty.

Intrachannel Crosstalk

Intrachannel crosstalk arises in transmission links due to reflections. This is usually not a major problem in point to point links since these reflections can be controlled. However, intrachannel crosstalk can be a major problem in networks. One source of this arises from cascading a wavelength demultiplexer (DEMUX) with a wavelength multiplexer (MUX) as shown in Figure-

Figure-Sources of intrachannel crosstalk. A cascaded wavelength demultiplexer and a multiplexer.

The DEMUX ideally separates the incoming wavelengths to different output fibers. In reality a portion of the signal at one wavelength say λi leaks into the adjacent channel λi+1 because of nonideal suppression within the DEMUX. When the wavelengths are combined again into a single fiber by the MUX, a small portion of the λi that leaked into the λi+1 channel will also leak back into the common fiber at the output. Although both signals contain the same data, they are not in phase with each other, due to different delays encountered by them. This causes intrachannel crosstalk. Intrachannel or homodyne crosstalk can be incoherent and coherent. Incoherent crosstalk occurs when the signal and interferer are from different optical sources. Coherent crosstalk occurs when the signal and interferer are from the same sources. The power penalty due to intrachannel crosstalk can be determined as follows.

Let P denote the average received signal power and εP the average received crosstalk power from a single other crosstalk channel. Assume that the signal and crosstalk are at the same optical wavelength. The electric field at the receiver can be written as

Here, ds(t)={0,1}depending on whether a 0 or 1 is being sent in the desired channel, ds(t)={0,1} depending on whether a 0 or 1 is being sent in the crosstalk channel, fc is the frequency of the optical carrier and are the random phases of the signal and crosstalk channels respectively. It is assumed that all channels have an ideal extinction ratio of ∞.

The photodetector produces a current that is proportional to the received power within its receiver bandwidth. This received power is given by

Assuming ε << 1, we can neglect the ε term compared to the term. Also the worst case above is when the cos(.) = – 1. Using this, we get the received power during a 1 bit as

and the power during a 0 bit as

Therefore, using (3.13) power penalty can be written as

Interchannel Crosstalk

 Interchannel crosstalk can arise from a variety of sources. A simple example is an optical filter or demultiplexer that selects one channel and imperfectly rejects the others, as shown in Figure-

Figure – Sources of interchannel crosstalk. An optical demultiplexer (DEMUX).

Estimating the power penalty due to interchannel crosstalk is fairly straightforward. If the wavelength spacing between the desired signal and the crosstalk signal is large compared to the receiver bandwidth, (3.15) can be written as

Therefore, in the worst case, we have


Using the above equation, power penalty can be written as-